Method for estimating time and frequency offset in an OFDM system

ABSTRACT

The synchronization technique invention uses inherent characteristics of the frequency domain representation of the data symbols. By computing a differential-in-frequency function across a large number of OFDM tones, robust estimates of time and frequency offset can be easily obtained. The technique also allows the system designer to directly trade performance in the presence of channel impairments against signal processing complexity. Analysis and simulation have shown good performance in the presence of noise and channel delay dispersion, impairments that are the harshest in a wireless environment. Prior techniques for OFDM synchronization have focussed on the time domain representation of the signal. Those that have recognized the translation of time and frequency offset to the frequency domain have not considered the systematic modification of the signal by the offsets.

This patent application is related to U.S. Ser. No. 09/128,738 filedAug. 5, 1998 by Alamouti, Stolarz, & Becker entitled “Vertical AntennaAdaptive Array”, and U.S. Ser. No. 08/796,584 by Alamouti et al.,entitled “Method for Frequency Division Duplex Communications,” assignedto AT&T Wireless Services and incorporated herein by reference.

FIELD OF THE INVENTION

The invention relates to a method to synchronize a multicarriertransmission system.

BACKGROUND OF THE INVENTION

Synchronization techniques for OFDM have been extensively studied:obtaining good performance under a variety of channel conditions withminimal signal processing is challenging. A good OFDM synchronizationtechnique will be applicable to more than the wireless high speed datacommunications system currently being studied—OFDM is being used orbeing considered in a variety of LEC networks in the form of ADSL, inDigital Audio Broadcast systems, in cable modems, and in digitaltelevision systems. OFDM is a special case of multicarrier transmissionsystems. The techniques described herein are generally applicable toother forms of multicarrier systems, e.g., discrete multitone (DMT)systems.

SUMMARY OF THE INVENTION

The invention uses inherent characteristics of the frequency domainrepresentation of the data symbols. By computing a differential infrequency function across a large number of OFDM tones, robust estimatesof time and frequency offset can be easily obtained. The technique alsoallows the system designer to directly trade performance in the presenceof channel impairments against signal processing complexity. Analysisand simulation have shown good performance in the presence of noise andchannel delay dispersion, impairments that are the harshest in awireless environment.

Prior techniques for OFDM synchronization have focussed on the timedomain representation of the signal. Those that have recognized thetranslation of time and frequency offset to the frequency domain haveapparently not considered the systematic modification of the signal bythe offsets.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 is a block diagram of OFDM transmitter.

FIG. 2 is an OFDM block structure.

FIG. 3 is a block diagram of OFDM receiver.

FIG. 4 depicts a plot of phase vs. tone number, no timing offset.

FIG. 5 depicts a plot of phase vs. tone number, 1 sample timing offset.

FIG. 6 shows differential in frequency constellation with 10 sampletiming offset.

FIG. 7 shows the 4^(th) power of differential in frequencyconstellation.

FIG. 8 shows the phase vs. tone number with 1/10^(th) tone frequencyoffset.

FIG. 9 shows the timing offset constellation with 6 dB SNR and 10 sampletiming offset.

FIG. 10 shows differential in frequency constellation with ½ tonefrequency offset and 22 samples timing offset.

FIG. 11 shows the effect of noise on individual tone phase.

DISCUSSION OF THE PREFERRED EMBODIMENT

I. Introduction

The problem of joint timing and frequency offset estimation is criticalfor the proper operation of a digital transmission system. It is also adifficult problem when it comes to system implementation. The problem iscompounded by the fact that timing offset may often create similarlooking signal impairments to frequency offset. A number of priorstudies have attempted to find timing and frequency offset estimationtechniques that are robust in the face of a wide variety of impairmentspresent on a wireless channel.

We focus on the problem of joint timing and frequency offset estimationfor an OFDM system. We first present a brief overview of an OFDM system.We review prior work in the area and describe our system's requirementsas an specific example of OFDM system design. We conclude by showing theperformance of the method in the presence of a variety of channelimpairments and describing practical limitations of the technique.

II. OFDM Overview

Delay spread, that is the time dispersion of a brief transmitted pulse,is a major impediment to high speed data transmission in the outdoors,high mobility wireless environment. Similarly, signal reflections in thelandline telephone copper cable plant or in a TV cable systems coaxialcable plant create similar dispersion and make high speed communicationsover the local access network difficult. In a wireless system, distantsignal reflectors can create several microseconds of delay spread. Thehigh mobility (i.e., high vehicle speeds) environment will create arapidly changing channel. Adaptive channel equalization techniques havebeen used to combat some of this effect, but are limited in theirability to deal with large amounts of rapidly changing delay spread,especially when the system is transmitting at the high symbol ratesneeded to attain high throughput. Conversely, using low symbol rates tomitigate channel dispersion will require dense constellations in singlecarrier systems. Dense signal constellations will be subject todegradation due to noise, interference, and fading.

In an OFDM system, a single higher speed bit stream is separated into amultiplicity of slower speed bit streams, each of the slower bit streamsused to modulate one of a set of carriers. The carriers are chosenwithin a single bandwidth and, for an OFDM system, with their inter-tonespacing chosen to insure orthogonality of the resulting combinedwaveforms. Generic multicarrier systems are not bound by the strictothogonality constraints but, for ease of implementation will often usevery similar techniques to the OFDM system described herein. Bydesigning the transmission in this way, the benefit of low symbol ratescan be attained without the penalty of dense constellations—eachindividual carrier has a sparse constellation, often QPSK. The totalcapacity of the OFDM system is determined by the union of a large numberof these individually sparse constellations. Typically, the OFDMtransmitter would be implemented by generating a series of complexnumbers, representing the phase of the individual tones, and using theInverse Discrete Fourier Transform (IDFT) to convert the series of tonesinto a time domain waveform. For practical reasons, in order to make theanalog reconstruction and antialiasing filters realizable, the size ofthe IDFT (or more often the Inverse Fast Fourier Transform (IFFT)) isusually larger than the number of tones. In addition, extra samples areprepended and appended to the transformed waveform. These “cyclicprefix” and “cyclic suffix” samples make the transmitted signal morerobust against time dispersion and timing offset—as long as the receivedsignal, plus its various delayed copies, are sampled within the cyclicextension, the constant amplitude of the received spectrum will bepreserved. At the transmitter, the summation of a large number ofsinusoids can generate a signal with a large peak-to-average waveform. Apeak control function is used to minimize this effect, relaxing therequirements on RF amplifier stages. FIG. 1 shows a simplified blockdiagram of an OFDM transmitter.

Modulated tones occupy only the first group of input tones to the IFFT(about one third of the total number of tones for the OFDM system here).The rest of the IFFT inputs are set to zero, insuring that the outputwaveform has no signal energy at higher frequencies and thusoversampling the resulting waveform. As stated above, this simplifiesthe requirement for antialiasing filtering of the final waveform.Besides this oversampling, other redundant information is added to theOFDM waveform—the IFFT waveform samples are cyclically extended bothbefore and after the desired set of samples. For an N point IFFT, thecyclically extended suffix samples are simply copied from the beginningof the waveform as (In the equations here, upper case letters indicatesignal representations in the frequency domain. Lower case lettersindicate the time domain):x _(i+N) =x _(i)

The cyclic prefix is computed similarly, copying the samples at the endof the waveform to the front of the extended waveform.

Two other segments that are added to the IFFT waveform are the windowedsection and the guard interval. The guard interval is a period duringwhich the transmitted samples are all zero. At the expense of a slightlyreduced transmission efficiency, the presence of this guard intervalhelps to insure that the samples received during a given OFDM block arenot contaminated with delayed samples from a previous block. Since theOFDM system described here will be used in packet mode, it is expectedthat the transmitter will be turned on and off during each OFDM burst.To prevent the radiated signal from “splattering” outside of itsassigned channel, “windowing” of the OFDM samples is used. By shapingthe transmitter ramp up and ramp down with a raised cosine pulse shape,the system bandwidth is confined to little more than the bandwidthoccupied by the set of OFDM tones alone. Windowing also reduces thesystem's sensitivity to frequency offset and doppler by reducing theamplitude of the interference contribution from adjacent tones. FIG. 2OFDM block structure illustrates these items and provides the particularparameters used for the system described here.

The OFDM receiver structure mirrors the operation of the OFDMtransmitter. While the technique described here is applicable to amulticast network, that is, a network with one transmitter and severalindependent receivers, to simplify the description a simple onetransmitter, one receiver network is presented. It can readily be seenthat each receiver in a multicast network could individually perform theoperations described on a common transmitted signal. In the receiver,extra samples are stripped from the received waveform and the resultingwaveform is transformed into the frequency domain. Here, assuming thatthere was no frequency or timing offset in the time domain waveform, aseries of tones are processed, with each conveying one of the lowerspeed bit streams. If QPSK were used at the OFDM transmitter to modulateeach tone, then each OFDM tone burst would convey 2 bits of informationper tone in the phase of that tone. A simplified OFDM receiver is shownin FIG. 3.

While the relatively long symbol period of the OFDM waveform makes thesystem resistant to delay spread, incorrect timing phase or frequencyoffset in the received signal will quickly degrade system performance.Most of the extra samples in the time domain waveform insure that theamplitude of the received OFDM spectrum remains constant in the presenceof delay spread and timing offset, due to the cyclic shifting propertyof the Fourier Transform. However, information is conveyed in the phaseof the received tones, so the damaging effects of timing offset must becorrected.

Frequency offset has a different effect on the received waveform. Whilethe OFDM transmitter generated orthogonal tones by creating a specificlinear harm onic relationship between all the tones, frequency offsetshifts each tone by an equal amount, creating an affine relationshipbetween tone frequencies, thus destroying the linear harmonic structureand creating inter-tone interference. Like inter-symbol interference ina single carrier system, frequency offset allows energy from adjacenttones to “bleed” into a desired tone, reducing system performance.

III. System Parameters

As an example of a practical system, the OFDM system here is intended toprovide peak end user data rates of up to 384 kb/s. As shown in FIG. 2,this system transmits OFDM bursts with a duration of 288.462 μsec. The512 FFT samples represent the transform of 189 discrete complex tones.These 189 tones are spaced every 4.232 kHz (skipping the three tonesnearest the center frequency), so the total bandwidth is 812.5 kHz. WithQPSK modulation on each complex tone, this means that the raw channelrate is 1.3104 Mb/s. With a rate ½ Reed-Solomon code plus framing andcontrol overhead, the peak end user capacity of 384 kb/s is easilyattained.

IV. Prior Work

Schmidtl's and Cox's papers in the IEEE Transactions on Communicationsand in the Proceedings of the International Communications Conferencedefine techniques for estimating time and frequency offset based onsignal processing in the time domain, using ‘pilot’ signals. While thepilot signals are useful for establishing a reference for calculatingsynchronization parameters, they require that transmit energy beexpended and signal bandwidth be consumed that could have been used totransmit end user information.

Moose's technique similarly requires repetition of an OFDM block,essentially using one of the repetitions as a ‘pilot’ signal. Frequencyoffset is computed by comparing samples of the first block to the secondin the time domain. This technique is only used for frequency offsetestimation. It does not address the need for time synchronization.

The work of Pollet, et. al., describes the degradation in systemperformance when adequate timing synchronization is not used.

Finally, van der Beek, et. al., describe a timing synchronizationtechnique that relies on inherent redundancy in the OFDM time domainwaveform. Because the cyclic extension of the FFT samples are simplyordered copies of other samples in the waveform, this technique relieson computing the time correlation between the repeated samples toestimate the timing offset. [J.-J. van de Beek, M. Sandell, M. Isaksson,and P. Borjesson, “Low-complex frame synchronization in OFDM systems,”in Proc. ICUPC, November 1995, pp. 982-986.] There are relatively fewcyclically extended samples in the OFDM block, so the robustness tonoise may be somewhat limited, processing a single OFDM block.

None of the previous techniques process the signals in the frequencydomain as the current scheme does, and none recognize the inherentstructure of the frequency domain signals that this scheme relies on.Except for van der Beek's scheme, all require the use of redundant pilotsignals.

V. Approach

Frequency offset causes each tone in the OFDM cluster to be shifted infrequency by the same amount. The complex OFDM time domain waveformappears to be rotated on a time-sample by time-sample basis by acontinuously rotating phasor. While this effect is instructive tounderstand how to undo the effect of frequency offset, there is nothingimmediately obvious about the time domain waveform that suggests how thefrequency offset can be estimated. Timing offset similarly shifts allsamples by a fixed time interval which, again by itself, gives littleinformation that allows estimation of the amount of offset whenobserving the time domain waveform. The current method performsestimation of the time and frequency error in the frequency domain,unlike previous approaches which did their estimation in the timedomain. In addition to realizing the other advantages described herein,a frequency domain approach to synchronization simplifies thearchitecture of the receiver.

Timing Offset Correction

First, consider the effect of timing offset on the OFDM tones. FIG. 4depicts a plot of the phase versus frequency (here, frequencycorresponds to FFT output bin number) for a randomly generated OFDM tonecluster with no timing offset. It can be seen that each tone takes onone of a discrete set of phases (one of four, in this case, since thetones are QPSK modulated). Each tone is independently modulated, so thetransitions between the phase of tone_(i) and tone_(i+1) are randommultiples of π/2. For this, and the subsequent phase plots, the FFT sizeis 512 and there are 189 active tones.

In contrast, FIG. 5 depicts a plot of phase versus frequency where theOFDM signal has been delayed by one time sample (462 μsec in the examplesystem). It can be seen that the tone modulation is still present,however, each tone has a phase that is slightly offset from the previousone. As shown in the section of this memo that presents the analysis ofthe effects of timing offset, this is exactly what is expected.

To understand how timing offset has changed the received signal, firstconsider an OFDM waveform x_(n), generated from a series of OFDM tones,X_(m):$x_{n} = {\frac{1}{\sqrt{N}} \cdot {\sum\limits_{m = 0}^{N - 1}{X_{m} \cdot {\mathbb{e}}^{j \cdot \frac{2 \cdot \pi \cdot n \cdot m}{N}}}}}$

At the OFDM receiver, the waveform is assumed to have a timing offset,Δt. By the shifting property of the Fourier Transform, a time delayedsignal in the time domainr(t)=x(t−Δt)

Has as its transform:R(f)=X(f)·e^(−j·2·π·f·Δt)

So, as demonstrated in FIG. 5, it can be seen that the time delay hascaused a phase rotation in the frequency domain that increases linearlywith frequency. Consider how the Discrete Fourier Transform of thesampled receive signal is affected by timing offset:$R_{n} = {X_{n} \cdot {\mathbb{e}}^{j \cdot \frac{{2 \cdot \pi \cdot n \cdot \Delta}\quad t}{N}}}$

For any adjacent pair of tones, R_(i) and R_(i+1),${\Delta\quad t} = {\frac{N}{2 \cdot \pi} \cdot ( {{\arg( R_{i} )} - {\arg( R_{i + 1} )}} )}$

This suggests an approach that could be used to estimate the amount oftiming offset: measure the differential phase from one tone to the nextand adjust the sampling point to compensate. It is instructive to notethat as shown in FIG. 5 and in the equation above, timing offset hascreated the same phase difference between every pair of tones: R_(i) andR_(i+1). This means that, while noise and other impairments may perturbthe individual tone phases, there is a systematic change in phase fromtone to tone due to timing offset. By estimating timing offset acrossmany or all the adjacent tone pairs in a burst, it is feasible toaccurately estimate the timing offset in the presence of a collection ofother impairments: frequency offset, noise, fading, delayspread/frequency selective fading, etc. Particular performance resultsare presented in the next section. Of course, the same inherentcharacteristics that cause each adjacent pair of tones to exhibit theidentical phase difference also cause each tone pair separated by Nother tones to exhibit a phase difference that is different than thatbetween adjacent tones, but also identical to all other N tone separatedtone pairs. Further, if it is necessary for the receiver to avoid usingcertain tones for phase estimation, it is possible to account for themissing tones by appropriately scaling the phase differences. Note—forthe purposes of this description, the terms ‘tone’ and ‘carrier’ areinterchangeable, the latter term being preferred for genericmulticarrier systems, the former used in the context of OFDM systems.

Two useful characteristics of the algorithm become apparent: (1) thatnoise effects may be partially or mostly mitigated by the tone-to-tonedifferential nature of the algorithm and (2) that the essential natureof the algorithm creates a weighting of timing estimates based on thelikely validity of each estimate. These observations are explained ingreater depth below.

One may think of the process of estimating timing offset as thedifferential detection of tone phase from one tone to its neighbor, infrequency. For QPSK, individual tone modulation can be removed from theestimate by raising the differentially detected phase to the 4^(th)power. Alternatively, each tone phase could be compared to thetransmitted signal constellation and the complex conjugate of thenearest signal used to rotate the signal to the positive real axis.While noise and other impairments will occasionally create incorrectdecisions about the transmitted signal and thus the received signalphase, these effects will average out across a number of estimates. Asanother alternative, if the correct transmitted data sequence were knownin advance by the receiver, this information could be used to determinethe expected phase of each tone. This information could be used asdescribed to rotate the received signal to the positive real axis. Togain the signal to noise advantage of averaging the tone-to-toneestimates, it is possible to average the phase of each measurement. ForN tones in an OFDM block, this approach requires the equivalent of N−1arctangent calculations, which is prohibitively expensive in a real-timesystem. A preferred embodiment, which gives better performance undermany channel conditions, is to separately average the in-phase andquadrature components of the processed constellation points andcalculate the phase of that signal to determine the timing offsetcorrection. FIG. 6 shows the constellation that results afterdifferential-in-frequency detection of the individual tones. FIG. 7shows the same constellation when each signal point is raised to the4^(th) power. Simulation results show that for reasonable amounts oftiming offset, the phase angle of the resultant signal is directlyproportional to the timing offset. The proportionality breaks down whenthe timing offset creates a very large rotation of the estimationsignal. As the estimation signal approaches the (−1,0) point in thesignal plane, there is an obvious 180 degree phase ambiguity, whichcould be addressed by processing the received signal instead of the4^(th) power signal. There is an additional degradation in the systemperformance as the timing offset causes the FFT waveform to wander faroutside the cyclic extension region into the windowed section of thedata. Neither of these limitations are considered serious, since thistiming estimation algorithm in intended to provide a steady statetracking control signal. Coarse initial timing acquisition (to withinreasonable fraction of the OFDM block) is sufficient to get this timingrecovery algorithm started.

Frequency Offset Correction

Having addressed timing offset, a technique to estimate frequency offsetis needed. FIG. 8 shows the OFDM signal phase versus frequency, thistime with a frequency offset of 1/10^(th) the tone spacing. While thediscrete QPSK phase levels are still evident, two effects can be seen:First, all the tones' phases are shifted uniformly on average. Second,there is a slow variation of phase across the OFDM spectrum, giving theappearance that the frequency offset is creating beat frequencies withthe individual tones. By averaging across a number of tones, the secondeffect can be removed, leaving the phase offset on each tone as apredictor of frequency offset. As it was for timing offset, while noiseand other impairments may affect individual tone phases, the systematicchange in tone phase is present for all tone pairs and can be combinedto obtain a robust estimate.

As before, assume that a series of samples, x_(n) are generated at thetransmitter. With a frequency offset of Δf the received samples, r_(n)are:$r_{n} = {x_{n} \cdot {\mathbb{e}}^{j \cdot \frac{{2 \cdot \pi \cdot n \cdot \Delta}\quad f}{N}}}$

-   -   or, combining this with the previous expression for the received        tones,        $R_{n} = {\frac{1}{\sqrt{N}} \cdot {\sum\limits_{m = 0}^{N - 1}{( {x_{m} \cdot {\mathbb{e}}^{j \cdot \frac{{2 \cdot \pi \cdot m \cdot \Delta}\quad f}{N}}} ) \cdot {\mathbb{e}}^{{- j} \cdot \frac{2 \cdot \pi \cdot n \cdot m}{N}}}}}$

As was used for estimating timing offset, frequency offset is similarlyestimated by calculating the average phase of the ensemble of thereceived tones. Again, the 4^(th) power signals may be used to removethe modulation. Alternatively, the other techniques described earliermay be employed to estimate the transmitter phase based on known orinferred transmit data. As was the case for timing offset, it ispossible to calculate either the phase of the average signal or theaverage of the phases of the individual signals. For moderate amounts offrequency offset (less than 70% of the tone spacing), either techniqueworks well. In fact, the lower complexity estimate (phase of the meansignal) tolerates larger frequency offsets well—more than 110% of thetone spacing. For typical OFDM systems, frequency offsets and Dopplershifts will generally be a fraction of the tone spacing.

Phase Offset Correction

Finally, it is necessary to consider the effect of phase offset in thechannel. The effect of the phase rotation can be expressed as:r _(n) =x _(n) ·e ^(j·Δφ)

So the received tones are given by:$R_{n} = {\frac{1}{\sqrt{N}} \cdot {\sum\limits_{m = 0}^{N - 1}{( {x_{n} \cdot {\mathbb{e}}^{j \cdot {\Delta\phi}}} ) \cdot {\mathbb{e}}^{{- j} \cdot \frac{2 \cdot \pi \cdot m \cdot n}{N}}}}}$

The constant rotation factor can be factored out of the summation,indicating that a constant phase rotation of the baseband waveformresults in a uniform rotation of each of the transformed tones.

A constant phase offset in the channel gives the same appearance to thereceived signal constellation that frequency offset produces. Bymeasuring the rotation of the constellation and driving that parameterto zero by modifying the baseband rotator's phase updates, bothfrequency offset as well as phase rotation of the channel will becompensated for. Actually, in terms of the final signal constellation,the phase rotation caused by frequency and phase offset will not be thelimiting factors—a system which uses differential (in time) detection ofthe signal phase is immune to fixed or (moderate amounts of) changingrotation. The most significant degradation caused by frequency offset isthe inter-tone interference caused by the loss of orthogonality, whichonly requires that the frequency offset be approximated—phase lock isnot essential.

VI. Simulation Results

To determine the utility of the timing and frequency offset estimationalgorithm, it is necessary to assess its performance in the presence ofa variety of channel impairments. In particular, it is necessary toconsider how the algorithm behaves in the presence of the followingimpairments:

-   -   White Gaussian noise    -   Frequency offset while making timing offset estimates    -   delay spread/frequency selective fading

These impairments were simulated using Mathcad both individually and invarious combinations. Results are presented below.

First, AWGN was simulated at varying signal to noise ratios. Even whenthe SNR was 6 dB, giving a 4^(th) power differential in frequencyconstellation as shown in FIG. 9, the timing offset estimator was ableto track the timing offset based on one OFDM block. Table 1 shows somerepresentative timing offset estimates at various SNRs and differenttiming offsets for the two measures (phase of mean and mean of phase).TABLE 1 SNR Timing offset Mean of Phase Phase of Mean (dB) (samples)Estimate (samples) Estimate (samples) 6  3 2.82 4.20 8  3 3.06 4.54 12 3 3.07 2.84 20  3 3.00 3.11 6 10 8.04 9.94 8 10 9.45 8.06 12 10 9.909.27 20 10 10.01 10.04 3 −22   −7.30 −19.20 8 −22   −20.7 −22.40 12−22   −22.06 −21.33 20 −22   −21.96 −22.00 6  40⁵ 24.01 42.37 8 40 20.5539.17 12 40 33.92 39.64 20 40 39.97 39.94

It can be seen from these AWGN results that, even in the presence ofpoor SNRs, the algorithm tracks timing offset well, preserving thedirection of adjustment and, even in the worst case, preserving thegeneral magnitude of the adjustment needed. Again, it must be noted thatthese results are for single OFDM block processing. Averaging estimatesover successive blocks improves the estimate in the presence of noise.At least in AWGN, simulation results show that the mean of the phaseestimate tracks more closely to the actual timing offset than the phaseof the mean estimate. While the former estimate is more costly, from asignal processing perspective, with fewer tones' phases differentiallycompared and with multi-block averaging, it may be preferable to usethis method under some circumstances.

Next, it is instructive to look at the effect that frequency offset hason timing offset estimation. As FIG. 8 suggests, frequency offset isunlikely to have much effect on the timing offset estimation. To verifythis, the same simulations were conducted as were presented in Table 1,this time with various amounts of frequency offset. Representativeresults are listed in Table 2. TABLE 2 Timing Frequency Offset Mean ofPhase Phase of Mean SNR offset (fraction of tone Estimate Estimate (dB)(samples) spacing) (samples) (samples) 8 10 .1 8.89 8.04 8 10 .5 7.829.05 8 10 1.0 7.01 13.86 20 −22 .1 −22.02 −21.93 20 −22 .5 −21.94 −22.3120 −22 1.0 −18.23 −23.53

While these numbers show the overall performance of the timing offsetestimator, it is instructive to examine the signal constellation thatthe estimator is working with. FIG. 10 shows the effect on thedifferential in frequency constellation due to a large amount ofsimultaneous timing and frequency offset. It can be seen that, while thetiming offset causes the constellation to “twist,” the frequency offsetcreates a variation in amplitude (due to inter-tone interference) thatcauses the constellation points to create loci of points originating at(−1,0). In spite of this amplitude variation, the centroids of thetwisted constellation points are essentially unchanged.

Finally, simulations of the timing offset estimator were run with delayspread. In one representative experiment, two equal rays with a spacingof 20.8 μsec (46 samples) were generated. With 10 samples of timingoffset, it would be expected that the estimator would indicate that 18samples of correction was needed¹. With a 10 dB SNR and 0.1 tonefrequency offset, the timing offset estimator indicated 15.7 samples forthe mean of the phase estimate and 19.8 samples for the phase of themean estimate. As shown by the real-time DSP results, delay spread hasminimal effect on the timing recovery algorithm.¹ With 10 samples of timing offset in the opposite direction to the 46samples delay spread, the centroid of the channel response would be(46-10)/2=18 samples from the initial sampling point.

The frequency offset estimator was next tried under simulated channelconditions. It is first worth noting, however, that it is not possibleto estimate the frequency offset in the presence of timing offset.Timing offset creates an ever-increasing rotation of the tone signalconstellation, making phase measurements meaningless. So, for theremaining simulations, it is assumed that either the timing offset hasbeen adjusted to zero by adjusting the sample clock or the individualtones have been rotated by the proper amount to compensate for thetiming offset.

VII. A Physical Interpretation of the Algorithms Performance

We have observed from the above that (1) the noise perturbation of thesignal phase does not create a significant problem for timing estimationand (2) the algorithm inherently provides a desirable weighting oftiming estimates. These ideas are expanded in this section to providesome intuition about why the timing offset algorithm behaves as well asit does.

First consider the effect of noise. In FIG. 11, observe an arbitrary setof tone phases. It is assumed that tone_(i) and tone_(i+2) provide goodestimates of the timing phase, but tone_(i+1) is a noisier signal,giving a poorer estimate of timing phase. Since the tone phases arebeing differentially compared to determine the overall timing phaseestimate, it can be seen that the positive bias in comparing tone_(i) totone_(i+1) will tend to cancel the negative bias in comparing tone_(i+1)to tone_(i+2). Thus, as the simulations revealed, good timing estimatescan be expected in rather poor SNR conditions. Further, since all of thetone phases have the same systematic bias due to timing offset, theoverall timing estimate will magnify this. Meanwhile, although analysissuggests that there is correlation between adjacent noise samples, incombination, the noise correlation across the entire band of frequenciesis small and does not seem to disrupt the timing estimate.

The wireless delay dispersive channel that this OFDM system must endurecreates frequency selective fading. From this, one might suspect thattones which are in the middle of a frequency null will have poor SNR andcannot be relied on to provide accurate phase estimates to drive thetiming algorithm. Worse, the attenuation of one tone generally alterstwo phase estimates—if tone_(i+1) is near an amplitude null, the phaseestimate derived from comparing tone_(i) to tone_(i+1) as well as thephase estimate obtained from tone_(i+1) and tone_(i+2) may both becorrupted. Experimental results have shown remarkable performance in thepresence of this frequency selective fading. Even when large clusters oftones are suppressed, the timing recovery algorithm performs well.Understanding why this was so provided additional insight into the powerof this algorithm.

As shown, each tone is used to estimate the phase of its neighbor. Thisis done by multiplying the complex conjugate of the first tone with thesecond tone. This process preserves amplitude information. By collectingthe sums of the real and imaginary parts separately, each differentialtone pair adds to the phase estimate with an imaginary componentproportional to the phase difference and to the amplitude. Further, eachdifferential tone pair contributes to the real component proportionallyto the amplitudes of the two tones. Thus, low amplitude, faded, and,probably noisy tones make a much smaller contribution to the overalltiming estimate than do high amplitude, most likely clean tones do. Theresult is that frequency selective fading (i.e., a time dispersivechannel) has little effect on the timing estimate. Further, since thisalgorithm raises the differential signal to the 4^(th) power to removeQPSK modulation, any amplitude differences between tone pairs will befurther accentuated.

VIII. Limitations and Practical Considerations

In the preceding description, it has been assumed that all of the OFDMtones would be used to do timing estimation. With 189 active tones, thisprovides 188 differential phase estimates to be averaged. The signalprocessing, on a per tone pair basis, is moderate, but cannot beneglected. First, to compute the product of a tone with its neighbor'scomplex conjugate requires 4 real multiplies and two real adds. Then, toraise this signal to the 4^(th) power requires 12 real multiples and 3adds, so the total number of operations is 16 multiplies and 5 adds. Itis likely that this can be improved slightly, but this still will be amoderate amount of signal processing. If all 188 tone pairs are used,the receiver will obviously be in the best position to get a good timingestimate, but it is instructive to see just how few tone pairs weresufficient.

In the presence of noise and other impairments, one tone pair is notsufficient to allow the receiver to properly estimate the proper timinginstant. With two or more tone pairs, the noise and interference effectsdescribed above are observed. As more and more tones pairs are used, thecontribution of each individual tone pair's phase estimate is lessenedso any applicable degradation of that estimate is likewise reduced.

In one embodiment of the invention, it is possible to average the phasedifference between each tone pair to arrive at an overall phase estimateand then to derive synchronization information from that average phase.In the preferred embodiment of the invention, the real and imaginarycomponents of the differentially detected difference between the tonesare each summed to arrive at the real and imaginary parts of an overallcorrection vector. The phase of this overall correction vector is thencalculated, representing the overall synchronization signal. Byperforming the calculations in this manner a simpler algorithm results,with one arctangent calculation required instead of one arctangentcalculation per tone pair.

The resulting invention provides a method for OFDM systems that iscapable of separately estimating time and frequency offset in thepresence of severe channel impairments, requiring no training sequencesor other overhead. The algorithm is capable of deriving high-qualityestimates on a single OFDM block or, with lower real-time overhead,could be used to track drift over a series of OFDM blocks.

Various illustrative examples of the invention have been described indetail. In addition, however, many modifications and changes can be madeto these examples without departing from the nature and spirit of theinvention.

1-82. (canceled)
 83. A method for receiving OFDM signals comprising thesteps of: applying signals received at input ports to a time-samplingelement; applying output signals of the time-sampling element to afrequency compensation element; applying output signals of the frequencycompensation element to a receiving section that includes an elementthat transforms signals related to said output signals of the frequencycompensation element to frequency domain to develop thereby frequencydomain signals; developing timing offset control signals from saidfrequency domain signals and applying the developed timing offsetcontrol signals to said time-sampling element; and developing frequencycompensation control signals from said frequency domain signals andapplying the developed frequency compensation control signals to saidfrequency compensation element.
 84. The method of claim 83 where saidstep of developing said timing offset control signals develops eachtiming offset control signal by averaging a plurality of timing offsetestimates that are computed from different pairs of said frequencydomain signals.
 85. The method of claim 84 where each of said timingoffset estimates corresponding to a pair of frequency domain signalscorresponds to a phase difference between said pair of frequency domainsignals.
 86. The method of claim 84 where said pairs are pairs ofsignals that are adjacent in to each other in the frequency domain. 87.The method of claim 86 where said each of said timing offset estimatesthat is developed from a pair of adjacent frequency domain signals isobtained by subtracting the phase of one of the signals in the pair,raised to a selected power, from the phase of the other of the signalsin the pair, raised to said selected power.
 88. The method of claim 83where said step of developing said frequency compensation controlsignals develops each frequency compensation control signal by averagingover an ensemble of said frequency domain signals.
 89. A method executedin a an OFDM receiver that includes a front-end section and a followingdetection-and-decoding section, where the detection-and-decoding sectiondevelops transformed signals that correspond to a transformation fromtime domain to frequency domain of signals provided by said front-endsection, the improvement comprising: developing timing offset controlsignals from said transformed signals and applying them to saidfront-end section for controlling time at which signals applied to saidreceiver are sampled; and developing frequency compensation controlsignals from said transformed signals and applying them to saidfront-end section for controlling